The present invention relates generally to high efficiency analog power amplifiers and, more specifically, to pulse-width-modulation ("PWM") type amplifiers and zero-voltage-switching multiresonant converters.
A typical PWM-type amplifier, i.e., the Class D amplifier, includes a comparator circuit coupled to the gates (or bases) of a pair of switching transistors that are coupled in series across a d.c. power source. The transistors are disposed in a conventional push-pull configuration. Reverse current by-passing or recovery diodes are also coupled in series across the d.c. power source, the junction of the diodes being coupled to the junction of the paired transistors. A low-pass filter is coupled to the junction of the paired transistors.
The comparator creates a rectangle-wave PWM signal from a modulating input signal and a triangle-wave carrier signal. The PWM signal is applied to the gates of the switching transistors, causing the transistors to be alternately switched on and off in accordance with the duration of the PWM pulses. The resulting demodulated signal passes through the low-pass filter and is output to a load.
Although highly efficient, conventional Class D amplifiers are subject to several drawbacks. The frequency of pulse width modulation in conventional Class D amplifiers is limited to a maximum of approximately 1 Mhz due to transistor switching losses. As a transistor is turned on and off rapidly, switching transients involving high levels of current and voltage occur, whereby high switching stresses and losses are imposed upon the switch. (A detailed discussion of such stresses and losses may be found in U.S. Pat. No. 4,720,668, issued to Lee et al.) As a result, Class D amplifiers are restricted to operate in a bandwidth of approximately 50 to 100 Khz. Such limitation can preclude the use of Class D amplifiers in high frequency applications since, for minimum distortion, the switching frequency of these amplifiers should be at least five times that of the highest frequency component of the modulating input signal. See, Grant, et al., Power Mosfets Theory and Applications, 314 (Wiley 1989).
Furthermore, Class D amplifiers generate output distortion when their switching transistors are not operating simultaneously (i.e., one turning off while the other is turning on). This differential switching error can result in "switch conduction overlap;" i.e., both transistors conducting simultaneously and thereby creating a virtual short circuit between the terminals of the d.c. power source. Overlapping conduction can result in very high peak currents flowing between terminals of the d.c. power source which can cause distortion as well as dissipation and device rating problems.
An additional cause of output distortion in conventional Class D amplifiers is pulse amplitude error (i.e., crossover distortion) over the analog cycle of the modulating input signal. As noted above, Class D amplifiers include a pair of switching transistors and recovery diodes. When an analog input signal passes from a positive to negative half cycle, effective output drive is transferred from one transistor and recovery diode to the other. This transition generates a crossover distortion component into the output waveform resulting from recovery diode overswings and forward voltage drops of the "on" transistor.
Finally, Class D amplifier output is also subject to high-frequency ripple distortion derived from the frequency of the carrier signal used to create PWM waveforms.
In addition to amplifiers, the technique of PWM has been applied to the field of power converters. A conventional PWM-type converter includes two switches; an active switch (i.e., a switching transistor) and a passive switch (i.e., a diode). To minimize switching stresses and losses experienced when switching between on and off states (and thereby maximize switching frequencies), resonant circuits have been incorporated into PWM-type converters to establish zero-current or zero-voltage conditions at a switch at the time of switching. Devices employing this technique include zero-current-switching quasi-resonant converters ("ZCS-QRC"), zero-voltage-switching quasi-resonant converters ("ZVS-QRC"), zero-voltage-switching multi-resonant converters ("ZVS-MRC") and constant-frequency zero-voltage-switching multi-resonant converters ("CF-ZVS-MRC").
ZCS-QRCs utilize an inductor/capacitor resonant tank circuit to force current passing through the switching transistor to oscillate, whereby the current is reduced to zero prior to turn-off. FIG. 1 illustrates a zero-current quasi-resonant switch 600. In contrast, ZVS-QRCs utilize a resonant circuit to shape the switching transistor's voltage waveform so that voltage levels reduce to zero prior to turn on. This enables ZVS-QRCs to operate up to and beyond frequencies of 10 Mhz. FIG. 2 illustrates a zero-voltage quasi-resonant switch 601.
Further, ZVS-MRCs utilize a resonant circuit that enables both the transistor and diode to operate with zero-voltage switching, thereby improving the performance of ZVS-QRCs. FIG. 3 illustrates a zero-voltage multi-resonant switch 602. ZVS-MRCs utilize the junction capacitances of all semiconductor devices to form a multi-resonant circuit. Finally, replacing the passive switch (i.e., diode) of a ZVS-MRC with a second active switch (i.e., transistor) enables output power to be controlled at a constant frequency, thereby creating CF-ZVS-MRCs. FIG. 4 illustrates a constant-frequency zero-voltage multi-resonant switch 603.
A conventional CF-ZVS-MRC circuit is illustrated in FIG. 5. Switch one (S1) operates with a constant switching frequency f.sub.s (i.e., f.sub.s =1/T.sub.s) and fixed on-time duration while switch two (S2) operates with constant frequency and variable on-time duration, thereby providing control of output power. To achieve zero-voltage turn on, S1 and S2 are switched on and off while their currents flow through antiparallel diodes D1 and D2. When S1 is off and S2 is on, C1 resonates with L. When S1 is on and S2 is off, C2 resonates with L. And when both S1 and S2 are off, all three resonant elements resonate with each other. As a result of this resonance, the voltage across each switch reaches zero prior to the switch being enabled (i.e., placed in an on state). In operation, the voltage across a switch remains at zero while the switch is enabled, and oscillates between a non-zero value and zero when the switch is disabled.
Additional detail regarding each of the foregoing converters, including the CF-ZVS-MRC, is provided in the following patents and publications, the disclosures of which are all hereby incorporated by reference: U.S. Pat. No. 4,720,668 issued to Lee et al.; M. M. Jovanovic, R. Farrington and F. C. Lee, "Constant-Frequency Multi-Resonant Converters," Virginia Power Electronics Center, The Bradley Department of Electrical Engineering, Virginia Polytechnic Institute & State University (undated) (hereinafter, "Jovanovic"), 56-65; D. Maksimovic and S. Cuk, "Constant-Frequency Control of Quasi-Resonant Converters," Technical Papers of the Fourth Annual High Frequency Power Conversion Conference, 241-253 (May 1989) (hereinafter, "Maksimovic"); R. Farrington, M. M. Jovanovic and F. C. Lee, "Constant-Frequency Zero-Voltage-Switched Multi-Resonant Converters: Analysis, Design, and Experimental Results," Power Electronics Specialists Conference Record, Vol. 1, 197-205, (June 1990), and W. A. Tabisz and F. C. Lee, "Zero-Voltage-Switching Multi-Resonant Technique--A Novel Approach to Improve Performance of High-Frequency Quasi-Resonant Converters," Power Electronics Specialists Conference Record, 14-22 (April 1988).